Power conversion apparatus

ABSTRACT

A power conversion apparatus for converting power from an alternating source to dc includes an input stage for receiving power from the alternating source, which includes an input filter, a rectifier for rectifying the alternating signal, a capacitor for storing energy from the rectified signal, and an output for outputting power from the rectifying means and the capacitor to the pulsed load. The pulsed load has at least one switched winding which receives power from the output, and wherein the capacitor is configured such that the voltage across the capacitor falls below 15% of the nominal peak rectified voltage of the source during each cycle of the alternating source. A converter of this kind provides benefits in that the current drawn from the ac supply falls within limits imposed by regulations and is simpler and cheaper than known converters of a similar power rating.

This invention relates to power conversion apparatus for use with, or aspart of, electrical apparatus which employs a pulsed current load. Theinvention is particularly applicable to, but not limited to, motors andpower supplies.

A large number of power electronics applications now require thegeneration of an intermediate dc voltage stage. Taking the example of avariable speed motor, shown in FIG. 1, the motor will derive a powersupply from a standard ac mains supply 10 at the local voltage andfrequency. The mains supply is fed to a mains filter 15, which serves toprotect the equipment from any spurious signals on the supply as well asto prevent unwanted signals generated by the equipment from beingpropagated over the supply. The ‘cleaned’ supply is then converted to dcby a dc link stage 20. The conversion to dc includes a bridge rectifierD1-D4 and some form of circuitry to produce a more even, dc-like, outputfrom the rectified signal, such as a capacitor. In this example, the dclink stage includes a boost Active Power Factor Correction stage (boostAPFC stage) 25 which will be described more fully below.

Another example of the use of an intermediate dc stage is inac-to-dc-to-dc converters which are used for dc power supplies. In thesetypes of power supply a mains ac supply is first converted to dc beforebeing converted to dc at the required voltage.

Typically, passive forms of power conversion which include anintermediate dc stage have a disadvantage in that they distort the shapeof the voltage and current waveforms drawn from the mains supply.Electromagnetic Compatibility Standards (EMC), such as those set out inBritish Standard EN 61000-3-2 (1995) and in the EMC Directive(89/336/EEC), define an acceptable level for the harmonic content in thecurrent which electrical equipment draws from a mains ac supply, as wellas an acceptable level of voltage distortion. These standards placeconstraints on how power conversion can be carried out. In addition, thepower factor is of concern since this will determine the rating ofcomponents such as the mains cable and whether the local mains supplysystem will be adequate.

The way in which the dc link is implemented varies according to therequired output power of the system. For a low power load, a dc outputcan be achieved very simply by placing a capacitor Cdc across the outputof the bridge rectifier, in parallel with the load. In order to maintaina highly regulated dc voltage, the dc side capacitor Cdc must have ahigh capacity. The large capacitor Cdc causes the input current to havea low power factor, and current is only drawn from the mains supply whenthe magnitude of the mains input voltage (Vac) is greater than the dcvoltage (Vdc). The input current resembles a series of spaced-apartpeaks, which causes a significant low frequency harmonic content. It isthis harmonic content that limits this approach to low power systemsonly, since for higher power loads the harmonic content would breach thelevels defined by the EMC regulations or lead to an unacceptably lowpower factor.

Various techniques have been developed to improve the quality of theinput current. Additional components can be added to the input filterstage, or the well known ‘valley fill’ circuit can be used. The valleyfill circuit improves the input current shape by splitting the dc linkcapacitor into two. For the standard bridge rectifier, current is drawnfrom the mains supply when the magnitude of the mains input voltage(Vac) is greater than the dc voltage (Vdc). However, for the valley fillcircuit, current is drawn when the magnitude of the mains supply isgreater than half of the dc voltage (Vdc/2). This means that current istaken from the mains for a longer period than that of the standardbridge rectifier, resulting in an improved power factor.

Due to the harmonic limitations of the above schemes, activelycontrolled input rectifiers are often used. The most common of these isthe boost APFC stage shown in FIG. 1.

Two control loops—a voltage control loop and a current controlloop—define the switching action of power transistor TR1. The voltagecontrol loop maintains the dc link voltage (Vdc) at the required level,and this is achieved by adjusting the amplitude of the current controlloop's current reference. The current control loop ensures the inputcurrent follows the reference defined by the voltage control loop. Thismulti-loop control structure dictates that one loop must be dominant.The general convention is that the current control loop dominates. Thishas the effect that dc voltage regulation (particularly during transientevents) is limited, due to the limited performance of the slave loop.Generally, increasing the value of the dc link capacitance (Cdc)compensates for this limitation.

FIGS. 2 and 3 show both the start-up transient and the steady stateperformance of the converter. Initially (0<t<0.005 seconds), theconverter is uncontrolled (the action of the boost stage is irrelevantif Vdc<|Vac|). Once the condition Vdc>|Vac| is achieved, the boost APFCstage actively controls the input current to be substantiallysinusoidal, with very good power factor. The high frequency superimposedon the main 50 Hz component is due to the switching action of the boostconverter and is directly related to the switching frequency of TR1. Theselected switching frequency for the converter must be sufficientlygreater than the harmonic limits imposed by the EMC standards.

The present invention seeks to provide an improved method of powerconversion and an improved type of power converter.

Accordingly, the present invention provides a power converter accordingto claim 1.

A converter of this kind has an advantage in that the current drawn froman ac supply can fall within limits imposed by EMC regulations, with asimpler and cheaper apparatus in comparison to known converters of asimilar power rating. For example, the link capacitor can be constructedas a film-type capacitor which is capable of coping with the requiredripple current and is cost-effective. The converter meets EMCregulations because the dominant frequency of the supply current, i.e.the frequency with the greatest amplitude, is equal to the frequency ofthe ac voltage supply and the majority of the harmonic content is at theswitching frequency of the pulsed current load and harmonics of thatswitching frequency. For a load which operates at a high switchingfrequency, such as a high speed motor or a switched mode power supply,the harmonic content will be located outside the frequency bands set outin the EMC standards.

Because the capacitor forming part of the dc link stage of the converterhas a small value, this has the advantages of reducing cost and physicalspace occupied by the converter. It is preferable that the size(capacity) of the capacitor in the dc link stage is matched to theamount of energy that is transferred from inductive elements in theinput filter and the load. Thus, when the load is in the form of amotor, when one of the motor windings (or winding pairs) is switchedoff, the energy stored in the winding is safely transferred to the dclink capacitor (or another winding) without creating an excessiveover-voltage event.

The converter is particularly well-suited to loads which can toleratesome variation in their received power and which operate at a switchingfrequency which lies outside the harmonic frequencies specified in theEMC standards. Switched, high speed motors such as switched reluctancemotors which drive a load such as an impeller are particularlywell-suited to being driven by a converter of this kind, since somevariation in the operating speed of the impeller can be tolerated.Surprisingly, the actual variation in the operating speed of an impellerhas been found to amount to less than 1% of the normal operating speeddue to the high inertia of a fast-moving rotor and impeller.

The impeller can form part of a fan or pump for moving a fluid, such asa gas or a liquid, along a flow duct. In the field of appliances, theimpeller can form part of a fan for drawing dirty air into a vacuumcleaner. In these types of application it is not critically importantthat the impeller always operates at a precise speed.

Embodiments of the invention will now be described with reference to thedrawings, in which:

FIG. 1 shows a known form of power converter for supplying power to amotor, using a boost APFC stage;

FIGS. 2 and 3 show the performance of the power converter of FIG. 1;

FIG. 4 shows a first operating state of the power converter of FIG. 1;

FIG. 5 shows the current drawn by the converter of FIG. 1;

FIG. 6 shows a second operating state of the power converter of FIG. 1;

FIG. 7 shows the variation in current between the first and secondoperating states of the power converter of FIG. 1;

FIG. 8 shows current waveforms for the power converter of FIG. 1;

FIG. 9 shows a third operating state of the power converter of FIG. 1;

FIG. 10 shows current flow in the winding of the motor shown in FIG. 1;

FIG. 11 shows power flows both into and out of the power converter ofFIG. 1;

FIG. 12 shows a first operating state of a power converter in accordancewith an embodiment of the present invention;

FIG. 13 shows voltage waveforms in the power converter of FIG. 12;

FIG. 14 shows a second operating state of the power converter shown inFIG. 12;

FIG. 15 shows a third operating state of the power converter shown inFIG. 12;

FIG. 16 shows current drawn from the supply by the power converter ofFIG. 12;

FIG. 17 shows a fourth operating state of the power converter shown inFIG. 12;

FIG. 18 shows current flows through the motor windings shown in FIG. 12;

FIG. 19 shows voltage across the dc capacitor shown in FIG. 12;

FIG. 20 shows the variation in voltage pulses supplied to the load shownin FIG. 12;

FIG. 21 a illustrates flux build-up in the load shown in FIG. 12;

FIG. 21 b illustrates the effect of reducing the conduction angle onflux build-up in the load shown in FIG. 12;

FIGS. 22-24 show the application of the power converter of FIG. 12 to avacuum cleaner;

FIG. 25 shows a known type of dc power supply;

FIG. 26 shows a dc power supply in accordance with an embodiment of thepresent invention.

FIGS. 27 and 28 are schematic sectional views, from the side and front,showing the application of the power converter of FIG. 12 to theagitator of a surface-treating device.

By way of comparison, and to provide a better understanding of thepresent invention, the conventional technique of active power factorcorrection will now be described in more detail with reference to FIGS.4-11.

Looking firstly at FIG. 4, the power factor correction circuit comprisesan inductor L2 and a power switching device, such as a power transistorTR1, placed in parallel across the output of the bridge rectifier D1-D4.A diode D5 and capacitor Cdc are placed in parallel with the powerswitching device TR1, with the dc output being taken across capacitorCdc.

FIG. 4 also shows a load in the form of a two-phase switched reluctancemotor. The first phase comprises a pair of power switching devices TR2,TR3 in series with a winding W1. The winding W1 forms one of the statorphase windings of the motor. A pair of diodes D6, D7 provide a path forthe ‘free-wheeling’ current through the winding when the switchingdevices TR2 and TR3 are switched off. A second phase of the motor hasthe same form as the first phase, comprising the power switching devicesTR4, TR5, winding W2 and diodes D8, D9. The operation of switch TR1 ofthe PFC circuit is independent of the operation of the motor switchesTR2 and TR3 (and TR4 and TR5). TR1 is controlled in a manner thatactively shapes the input current whereas TR2, TR3 are controlledaccording to the required acceleration or steady state running of themotor.

For simplicity, in the following description certain assumptions havebeen made:

the voltage across the dc link capacitor (VCdc) is constant and greaterthan the peak rectified voltage;

the switching frequency of TR1 is much greater than the switchingfrequency of the load (i.e. the switching frequency of TR1 is greaterthan the switching frequency of TR2-TR5).

Three states of operation are shown in FIGS. 4-11.

State 1—FIG. 4

The PFC switch TR1 is on and switches TR2, TR3 are off. The periodduring which TR1 is switched on is chosen so as to actively shape theinput current. Current flows from the ac supply, through the bridgerectifier D1-D4, inductor L2 and TR1. The on/off time of TR1 is chosenso that the current through inductor L2 (and thus the input current IL2)has the shape shown in FIG. 5.

State 2—FIG. 6

TR1 is off while TR2 and TR3 are on.

There are two current loops:

I1: With TR1 off, energy stored in L2 is transferred to Cdc, whichresults in a net reduction in the current in L2 as shown in FIG. 7.

I2: In the second loop, energy stored on Cdc is released through windingW1.

The net current flowing into Cdc is I1-I2. The average currents over aperiod of time are shown in FIG. 8. It can be seen that capacitor Cdcmust, at any time, make up the difference between input current IL2 andthe output current (IW1+IW2). This causes a voltage ripple on Cdc of theform shown in FIG. 8. The maximum ripple is ΔV. The size of ΔV isinversely related to the capacitance of Cdc, i.e. a small voltage rippleΔV requires a large capacitance.

State 3—FIG. 9

TR1 is off while TR2 and TR3 are off.

There are two current loops:

I1—With TR1 off, energy stored in L2 is transferred to Cdc.

I2—With TR2 and TR3 off, the current in winding W1 reduces and isrecovered back to Cdc.

While they are not shown, the current flows for winding W2 are the sameas for winding W1.

It should be clear from the above that while the overall input powerP_(IN), i.e. power taken from the ac supply, is the same as the overalloutput power P_(OUT), i.e. power delivered to the load, over one cycleof the mains supply, the input power profile is very different to theoutput power profile, as shown in FIG. 11. Capacitor Cdc copes with theinstantaneous difference between input power and output power. For ahigh power load, this demands that Cdc must have a large value. As anexample, for a 1.5 kW load, Cdc must have a value of around 200 μF.

In summary, this scheme provides a good, stable, output voltage Vdc andthe shape of the input current drawn from the supply is compatible withEMC standards, i.e. the dominant frequency component is the samefrequency as the ac mains frequency with the much higher switchingfrequency of switch TR1 superimposed on the 50 Hz signal. Input currentrises as TR1 is turned on and falls as TR1 is turned off. The penaltiesof this scheme are that the capacitor Cdc must have a large value,requiring a capacitor which is both physically large and expensive.

Small DC Capacitor Scheme

With the scheme according to the invention, as shown in FIG. 12, themains filter (C1, C2, L1) and bridge rectifier (D1-D4) are retained.However, in place of the inductor L2, switch TR1, diode D5 and largecapacitor Cdc, there is now only a single link capacitor Cdc. The linkcapacitor Cdc has a capacitance which is of a considerably smaller valuethan that of the larger capacitor Cdc shown in FIGS. 1-11. The sametwo-phase motor serves as the load, as before.

In overview, this scheme has the effect that, each time one of the motorphases is energised, the energy stored in the link capacitor Cdc israpidly removed to the point where the rectifier diodes D1-D4 begin toconduct and the required motor power is taken directly from the mainssupply. The continuous pulsing of power directly from the mains supplyto the motor windings W1, W2 results in a similarly pulsed input currentwaveshape, shown in FIG. 16. The input ‘π’ filter formed by C1, C2 andL1 reduces the peak input current to an acceptable level and introducesa continuous current wave-shape similar to that for an activelycontrolled boost APFC stage. The resulting currents in the windings W1and W2 are shown in FIG. 18.

Operation of the circuit will now be described in more detail. Fourstates of the circuit will be described.

State 1—FIG. 12

TR2 and TR3 are switched on to energise the winding W1.

Just before TR2 and TR3 are turned on, the voltage across Cdc is equalto the mains peak voltage, minus the voltage across two of the bridgerectifier diodes. As TR2 and TR3 are turned on, the voltage across Cdcfalls very quickly to the instantaneous value of the rectified mainssupply, as shown in FIG. 13. The voltage across Cdc falls very quicklybecause of the small capacitance of Cdc.

State 2—FIG. 14

TR2 and TR3 remain switched on to energise the winding W1.

When VCdc falls to the rectified voltage level, the current/powersupplied to the load is no longer supplied only by the capacitor Cdc butis also drawn directly from the mains supply, as shown by the currentflow in FIG. 14. Because Cdc stores very little energy, VCdc is forcedto follow the rectified input voltage. This results in a voltage rippleon Cdc of around 85-100%.

Power flow to the load (motor windings) is dominated by flow from themains supply to the windings. There is no significant intermediateenergy storage, as in the boost APFC stage previously described.

State 3—FIG. 15

TR2 and TR3 are switched off.

There are two current flows:

I1—C1, C2 and L1 form an input filter which reduces the switchingfrequency (motor) component of the input current. When TR2 and TR3 areturned off, current continues to flow in L1.

I2—After TR2 and TR3 have been turned off, current continues to flowthrough winding W1 and is recovered to Cdc.

The size of capacitor Cdc is heavily dependent upon the total energytransferred from winding W1 and from the inductor L1 forming part of theinput filter during the time that T2 and TR3 are switched off. It isalso heavily dependent upon the total energy transferred from winding W2and from the inductor L1 during the time that TR4 and TR5 are switchedoff. The capacitance is selected so that the maximum voltage appliedacross the capacitor Cdc is kept within a predetermined limit: in theembodiment described, that limit is selected to be 400-500V.

State 4—FIG. 17

TR2 and TR3 are switched off.

Here, all of the energy stored in the winding has been recovered andhence the winding current has fallen to zero. Current still flows intothe inductor of the input filter L1 and charges Cdc.

FIG. 16 shows the input current drawn from the ac supply. It can be seenthat the input current has a significant component at the frequency ofthe mains supply, and is modulated at the switching frequency of theload. The input filter (C1, C2, L1) restricts the size of the componentat the switching frequency, and it is preferable to match the inputfilter to the switching frequency. The provision of the small dc linkcapacitor Cdc allows the current drawn by the load closely to follow themains supply. The size of the dc link capacitor Cdc is chosen inaccordance with the work demanded by the load applied to the dc link. Asdescribed above, for a load in the form of a pulsed motor winding, thedc link capacitor Cdc must be large enough to accept all of the energytransferred from de-energised phase windings without exceeding thevoltage capability of the components, as shown in FIG. 19.

It is acknowledged that this circuit arrangement is not suitable for alltypes of load. Firstly, the large (near 100%) ripple component on the dclink voltage causes a significant variation, over the course of onecycle of the supply, of the power supplied to the load. When the load isa motor, this has the effect that the speed of the motor will vary aboutan average value at a frequency equal to twice the frequency of themains supply. Secondly, current pulses, at the switching frequency ofthe load, appear in the input current. This demands that the switchingfrequency of the load must be sufficiently high to lie outside thestrictly regulated bands set out in the EMC standards. However, even inview of the above, this circuit arrangement is well-suited to many typesof pulsed loads, such as a motor where the switching frequency is highand where it is acceptable for the speed of operation to vary with themains frequency. The load should have a high operating frequency, of theorder of 2 KHz or more, in order to comply with current EMCrequirements, which makes this arrangement best suited to high speedmotors, such as those operating at speeds in excess of approximately35,000 rpm. Surprisingly, it has been found that the variation in inputpower does not have a significant effect on the speed of the motor.Indeed, for a motor operating at 95,000 rpm, a peak-to-peak variation of800 rpm has been observed.

A number of other changes have been found to be required for optimumoperation of the new converter with a pulsed current load.

It is preferable to avoid any significant build-up of flux in the motorwindings. To avoid flux build-up in any magnetic material, thevolt-seconds applied during de-energisation must be substantially equalto the volt-seconds applied during energisation. For equal energisationand de-energisation periods, the flux build-up will be proportional tothe voltage applied.

FIG. 20 illustrates the sequence of voltage pulses which are applied tothe windings of the motor during one half cycle of the input supply. Dueto the small value of Cdc, the input voltage varies widely during thehalf cycle. During 0<Time<0.005 s, the amplitude of the voltage pulseduring the off period is greater than the amplitude of the voltage pulseduring the immediately preceding on period and, as a result, fluxbuild-up in the motor does not occur. However, during the period 0.005s<Time<0.01 s the amplitude of the voltage pulse during the off periodis less than the amplitude of the voltage pulse during the immediatelypreceding on period and, as a result, flux build-up will occur for equalperiods of energisation and de-energisation. FIG. 21 a illustrates howflux build-up can occur when the off period has the same duration as theon period.

We have found that the problem of flux build—up in the motor illustratedin FIGS. 12-26 can be avoided by reducing the conduction angle, i.e. theduration of the energisation period or ‘on’ pulse. FIG. 21 b illustrateshow flux build-up can be avoided in this way.

There are other factors which must be considered before the energisationperiod is reduced. Excessive reduction of the energisation period willresult in periods of no motor current, which will have a detrimentaleffect on the harmonic content of the input current drawn from thesupply. Also, there will be a need to increase the peak current if themotor is to develop the same rated output power with a reducedenergisation period.

A compromise has been found where the energisation period is reducedonly to the point where the problem of flux build-up is eliminated. Inthe embodiment of a high speed motor, we have found that acceptableresults can be achieved by reducing the conduction angle from 90° to82°. Of course, the conduction angle will differ for other applications.

The value of the dc link capacitor Cdc is only governed by therequirement to absorb recovered energy from the motor, particularlyduring motor acceleration. During normal operation of the motor, when aphase winding is de-energised the energy stored in that winding is fedback to the dc link capacitor Cdc. This recovered energy can be as highas 33% of the rated power of the motor. As a result of absorbing therecovered energy from the winding, the capacitor voltage increases.Sizing of the dc link capacitor Cdc must take this voltage rise intoaccount, to ensure none of the components connected to the dc linkcapacitor Cdc suffer over-voltage events. It will be appreciated thatpower electronic components are sensitive to over-voltage events.

FIGS. 22-24 show the application of the power converter to driving animpeller of a suction fan in a vacuum cleaner. The vacuum cleaner shownhere is an upright type of vacuum cleaner but the vacuum cleaner couldequally be a cylinder type of vacuum cleaner. As shown in FIG. 22, thevacuum cleaner 100 comprises an upstanding main body 110 with a fan andmotor casing 120 at its lower end for housing a motor and fan unit. Acleaner head 115 is mounted in a freely articulated fashion on the motorcasing 120. A suction inlet 116 is provided in the cleaner head 115through which dirt and dust can be drawn from a floor surface. The mainbody 110 supports separating apparatus 112 in the form of a cyclonicseparator which can separate dirt, dust and other debris from a dirtyairflow drawn in through the inlet 116.

The fan and motor housing 120 supports an impeller 130 and a motor todrive the impeller 130. In use, the motor rotates the impeller 130 at avery high speed (of more than 70,000 rpm) to draw air along the pathsA-H through the cleaner 100. Dirt-laden air is drawn into the cleanerhead 115 via the dirty air inlet 116. The dirt-laden air is carried byducting to a separator 112 which serves to separate dirt, dust and otherdebris from the air flow (path B). The separator 112 can be a cyclonicseparator, as shown here, or some other form of separator, such as afilter bag. Cleaned air leaves the separator 112 along paths C, D beforeentering, via path E, the fan and motor housing 120. A pre-motor filteris usually placed in the airflow path before the impeller 130 to filterany fine dust particles which were not removed by separator 112.

FIGS. 23 and 24 show the impeller 130 and motor which are housed in themotor housing 120. Sets of bearings 143 support a shaft 142 which isrotatable about an axis 146. The impeller 130 is coaxially mounted onthe shaft 142 at the upstream end of the shaft 142. Blades extendradially outwardly from the main body of the impeller 130 towards thehousing 135 within a channel 148 and, in use, serve to draw air into thehousing 135 in the direction shown. Shaft 142 is driven by the motorwhich, in this embodiment, is a switched reluctance motor. The motor hasa stator 140 and a rotor 150 which is rotatably mounted within thestator 140. FIG. 24 is a section through the motor along X-X′ of FIG.23. The motor is a two pole, two-phase switched reluctance motor. Itcomprises a stator 140 having four salient poles 140 a, 140 b, 140 c and140 d. Each pole 140 a-140 d has a number of turns of insulated wirewound around it. The turns on opposing pairs of poles are joined inseries to form one winding, e.g. the turns on poles 140 a, 140 b formwinding W1 shown in FIG. 12 and the turns on poles 140 c, 140 d formwinding W2 shown in FIG. 12.

The circuit shown in FIG. 12 is used to power and drive the motor. Acontrol system 160 is also provided. The shaft 142 has a sensor 155 fordetecting the angular position of the rotor 150. In use, the controlsystem 160 uses the information from the sensor 155, together with otherinformation, to energise sequentially the windings W1 and W2 and henceto cause the rotor 150 and the impeller 130 to rotate about the axis146, drawing air into the housing 135 along path F and exhausting airalong path G. The windings W1, W2 are energised by turning TR2-TR5 onand off in the manner previously described. Control systems of this kindare well known and do not need to be described further.

For a two-phase switched reluctance motor with a normal operating speedof around 95,000 rpm, we have found that the following component values,for the circuit shown in FIG. 12, provide good results:

C1=C2=220 nF;

L1=330 μH

Cdc=6.6 μF

The motor illustrated in FIGS. 23 and 24 has a small number of poles anda high operating speed. The invention is equally applicable to otherloads having a high switching frequency, such as a motor having a largenumber of poles and a low operating speed. An example of such a load isa surface-treating device, such as an agitator, in a domestic appliance.FIGS. 27 and 28 illustrate such an agitator in the form of a brush bar170.

The brush bar 170 comprises an elongated cylindrical sleeve 171 havingradially extending bristles on its outer surface, as indicated at 172.The brush bar is rotatably mounted on an internal coaxial shaft 173 viabearings 174, 175. The motor is mounted centrally within the brush barand comprises a stator 176 and a rotor 177. The rotor 177 is coaxialwith the stator 176 and surrounds it such that the rotor rotates aroundthe stator. The shaft 173 is fixed with respect to the stator 176 andthe brush bar 170 is arranged to rotate with the rotor 177. The motor isan eighteen-pole, two-phase switched reluctance motor. A winding for themotor is indicated at 178 in FIG. 28. In use, a control circuit, such asthat shown in FIG. 12, is used to power and drive the motor. Eachwinding is energised in dependence on information from an angularposition sensor (not shown) associated with the rotor.

The motor causes the brushbar 170 to rotate at a typical operating speedof 3,500 rpm. The brushbar 170 may be included in the vacuum cleaner 100of FIG. 22. The brushbar is mountable in the cleaner head 115, adjacentthe suction inlet 116. Rotation of the brush bar 170 causes the bristles172 to sweep along the surface to be cleaned, for example a carpet,agitating the fibres of the carpet to loosen dirt and dust and pickingup debris. The suction of air lifts the dirt and dust from the carpetand into the dirty air inlet 116, and hence into the dust separationchamber 112 of the vacuum cleaner. The brushbar 170 may also be includedin a floor tool for a vacuum cleaner.

DC Power Supply

A second application of the power converter is in a dc power supply. Atypical dc power supply for power ratings in excess of 1-2 kW is a fullbridge dc-dc converter, as shown in FIG. 25. At the mains supply side,there is an input filter 300 (L1, C1, C2) and a bridge rectifier 305.Due to the high power rating, a boost APFC stage 310 is usuallyincorporated next to ensure satisfactory input current harmonics. Byincorporating the boost APFC stage, Vdc_A will be maintained at a nearconstant dc voltage. The boost APFC stage is followed by a full bridgeconverter 315. With a constant dc link voltage Vdc_A, control of thefull bridge converter is straightforward, depending only on thevariation in load. The output of the fully controlled bridge 315 is fedto a transformer 320 and an output filter which includes an inductor L2and an output dc capacitor C4. Vdc_B is the dc output voltage of the dcpower supply. The switching frequency of the bridge converter 315 isselected to minimise the size of the output filtering components (L2,C4) whilst maintaining acceptable losses in the power electronic devicesof the bridge converter 315. However, the selection of the outputcapacitor C4 is further complicated by standard requirements that theoutput voltage should be ‘held up’ for a defined period after the inputsupply has been removed, i.e. the output should remain on for a fixedtime period after the input supply has been removed, such as during apower cut. This generally results in the capacitor C4 having a fairlylarge value, often in the range of 100 s of mF. Using a boost APFC stage310 has the same problems as in the power converter shown previously inFIG. 1, in that it requires C3 to be large (100-150 μF) and increasesthe component count, size and cost of the overall power supply.

Using a technique similar to that described previously, the power supplycan be modified in a way that removes the boost APFC stage 310,retaining only a capacitor C3 of significantly smaller value, as shownin FIG. 26. As a consequence of removing the boost APFC stage 310, Vdc_Anow has near 100% ripple. Power transfer from the bridge converter 315,through the transformer 320 to the output stage, which is a function ofthe dc link voltage (Vdc_A), now varies over time. The input current tothe transformer is now-taken directly from the mains supply, since thesmall capacitor C3 stores very little energy. As before, flux build upin the transformer must be avoided by imposing limits on theenergisation period of the transformer. The size of the small capacitorC3 is heavily dependent upon the total energy transferred from theprimary winding Np of the transformer and from the inductor L1 formingpart of the input filter during the operation of the bridge converter315.

Removing the boost APFC stage 310 has the apparent drawback that theswitching frequency of the bridge converter no longer defines the valuesof the output filtering components (L2, C4). Capacitor C4 now has to besized to cope with the varying power transfer, which is a function ofthe mains supply frequency. However, it has been found that the value ofcapacitance C4 which is required with this new scheme is similar to thatwhich would have been required previously, as the standard requirementfor the output ‘hold up’ period already dictates a large value ofcapacitor C4. The majority of the energy storage capacitance is presenton the low voltage side, which has advantages in both cost and size.

It will be appreciated that the invention is not limited to theembodiment illustrated in the drawings. Specifically, the invention canbe applied to multi-phase systems, for example with independentrectification for each phase.

1. A power conversion apparatus for converting power from an alternatingsource to dc, comprising: an input stage for receiving power from thealternating source, which input stage includes an input filter, arectifier for rectifying the alternating signal, a capacitor for storingenergy from the rectified signal, an output for outputting power fromthe rectifier and the capacitor to the pulsed load, wherein the pulsedload has at least one switched winding which receives power from theoutput, and wherein the capacitor is configured such that the voltageacross the capacitor falls below 15% of the nominal peak rectifiedvoltage of the source during each cycle of the alternating source.
 2. Apower conversion apparatus according to claim 1, wherein the capacitoris configured such that the voltage across the capacitor falls below 10%of the nominal peak rectified voltage of the source during each cycle ofthe alternating source.
 3. A power conversion apparatus according toclaim 1 or 2, wherein the capacitor is configured such that the voltageacross the capacitor falls below 5% of the nominal peak rectifiedvoltage of the source during each cycle of the alternating source.
 4. Apower conversion apparatus according to claim 1 or 2, wherein thecapacitor is configured to store the amount of energy which is releasedfrom the winding when the winding is switched off.
 5. A power conversionapparatus according to claim 1 or 2, wherein the pulsed load has aswitching frequency which is greater than 2 KHz.
 6. (canceled)
 7. Anelectrical apparatus comprising a power conversion apparatus accordingto claim 1 or 2 and a pulsed load.
 8. An electrical apparatus accordingto claim 7, wherein the pulsed load is an inductive load which isrepeatedly switched between an on state and an off state, wherein theduration of the on state is less than the off state so as to minimize oravoid flux build up in the inductive load.
 9. An electrical apparatusaccording to claim 7, wherein the pulsed load comprises a motor havingat least one switched phase winding.
 10. An electrical apparatusaccording to claim 9, wherein the motor is a switched reluctance motor.11. An electrical apparatus according to claim 9, further comprising animpeller which is driven by the motor.
 12. A vacuum cleaner comprisingthe electrical apparatus according to claim 11 and an airflow pathformed within the vacuum cleaner, wherein the impeller is a suction fanfor drawing air along the airflow path.
 13. An electrical apparatusaccording to claim 9, further comprising a surface-treating device whichis driven by the motor.
 14. An electrical apparatus according to claim13, in which the surface-treating device comprises an agitator which isrotatable by the motor.
 15. A vacuum cleaner comprising the electricalapparatus according to claim 14 and an airflow path formed within thevacuum cleaner, wherein the agitator is located in a cleaner head orfloor tool of the vacuum cleaner.
 16. An electrical apparatus accordingto claim 7, wherein the pulsed load is a power supply, and the switchedwinding comprises a transformer.
 17. (canceled)
 18. An electricalapparatus according to claim 8, wherein the pulsed load comprises amotor having at least one switched phase winding.
 19. An electricalapparatus according to claim 18, wherein the motor is a switchedreluctance motor.
 20. An electrical apparatus according to claim 19,further comprising an impeller which is driven by the motor.
 21. Anelectrical apparatus according to claim 10, further comprising asurface-treating device which is driven by the motor.
 22. A vacuumcleaner comprising the electrical apparatus according to claim 14 and anairflow path formed within the vacuum cleaner, wherein the agitator islocated in a cleaner head or floor tool of the vacuum cleaner and themotor is a switched reluctance motor.